MOSFET Switching Results
are two main parameters which need to be addressed when considering the efficiency
of a switching device, these are the switching losses and the conduction losses.
The switching losses arise during the turn-on and turn-off portions of the conduction
waveform and are calculated by integrating the power dissipation (I*V) over the
duration of these transient events. The conduction loss is simply the dissipation
during the conduction part of the waveform and is also calculated by integrating
the I*V function. Given the correct gate drive conditions mosfets can easily switch
from the off state to fully on in less than a microsecond. Since the full conduction
time in low power coilguns is of the order of 1-15ms we really only need to concern
ourselves with the conduction losses. The conduction waveform for coil D as shown
in fig 1.
1. Mosfet conduction waveforms.
first thing we can see from this data is that the switching time of the mosfet
is extremely fast compared to the total conduction time. The transition is so
fast that it requires the waveform to be captured with a microsecond timebase.
When the device is off the D-S voltage is sitting at the source level of 32V.
When the trigger pulse turns the device on the DS voltage collapses to 0V in much
less than 1ms and only begins to rise as current is conducted through the device.
In a similar way the DS voltage rapidly rises as the device is turned off. Notice
that the DS voltage waveform follows the shape of the current trace. This characteristic
shows the resistive nature of the mosfet. If we calculate the resistance of the
mosfet module based on the peak current and corresponding voltage we arrive at
a value of 13m.
This is consistent with the parallel combination of three BUZ11 devices which
each have a maximum RDS ON value of 40m.
Something to bear in mind is that since the DS voltage measurements are made at
the mosfet module terminals there will be some small voltage drop across the conduction
paths inside the module. The RDS ON values will for the devices will certainly
be less than the maximum specified on the data sheet.
can examine a typical G-S and DS turn off voltage waveform in a little more detail
as shown in fig 2.
2. Turn off voltage waveforms for gate and drain terminals.
at the G-S voltage we can see that it takes the opamp output about 18us to reduce
the gate voltage to zero. Calculating the gradient of this line gives a slew rate
of 0.53 V/us. Although the 358N data sheet doesn't specify an open loop slew rate,
this value is similar to the slew rate recorded in the data sheet voltage follower
pulse response graph. A slew rate of 0.5V/us is typical of general purpose
opamps but something faster will be needed for high power coilguns - there are
several options for fast gate drive circuitry which will be investigated in the
we look at the D-S voltage waveform we can see that the turn off transition begins
at 26us and is complete at 32us - the end of the avalanche breakdown (see below)
and the beginning of the oscillation. Now, the G-S voltage at 26us is just under
4V which is 1V greater than the 'typical' threshold voltage of 3V. The reason
why the gate voltage can drop so low before the D-S voltage begins to rise is
due to the low conduction current in each of the BUZ11 devices. In this coil A
example the current at turn off is 19A so the current per device is only 6.3A.
Data sheets sometimes contain a set of output characteristic curves which
show the drain current as a function of D-S voltage for discrete values of G-S
voltage. Looking at the Fairchild Semiconductor data sheet for the BUZ11 we find
that the maximum drain current at a G-S voltage of 4V is 2.5A which is much less
than the 6.3A calculated above.
reasonable to ask why this difference exists and there are two explanations which
come to mind. The first is that the devices might have lower threshold voltage
than the typical value. The second is that the G-S voltage is being measured at
the terminals of the MOSFET module which
contains some 'protection' circuitry between the module gate post and the mosfet
gate terminals. This means that, due to the gate capacitance, there could be a
significant voltage drop across this resistor/zener network as the charge flows
from the gate.
the first possibility yielded a threshold voltage of about 3.2V so this explanation
was abandoned leaving the protection network as the likely culprit. Some rough
calculations using the gate charge and slew rate indicated that there would be
about 0.5V dropped across the network. This would mean that, at the turn off transition,
the actual voltage at the mosfet gate would be about 4.5V with a corresponding
current of 6A per device and a total current of 18A. This is in much better agreement
with the actual current level. It might be worthwhile to investigate this further
from a measurement standpoint. This is an important lesson in realising that you
might not be measuring exactly what you think you're measuring. This will make
me pay more attention to the measurement layout in future.
Module BUZ11 RIP.
The threshold voltage measurement involved determining when a 'significant' current
was conducted through the module. Unfortunately this current became too significant
and coupled with the 32V across the D-S was enough to cook the BUZ11s before I
realised what was happening. Replacing the mosfets only requires a bit of desoldering
and this is actually a good opportunity to substitute the BUZ11s for BUZ22s. These
devices are rated at 100V so I'll be able to use the module for some 'intermediate'
voltage capacitor experiments.
device attribute of interest is the ability to 'absorb' voltage transients. Most
modern mosfets are designed to cope with a limited degree of avalanche breakdown
through the D-S path, these are described as 'avalanche rugged' or 'avalanche
rated' devices. The limiting factor governing this breakdown is the junction temperature
rise due to energy dissipation. Using this ability it's actually possible to drive
inductive loads such as solenoids without the need for a clamping or commutating
diode. Unfortunately this is not the case in a coilgun simply because the amount
of energy stored in the field at turn off is too large to be dissipated during
avalanche breakdown - the device junction would be cooked.
avalanche breakdown capability is, however, still useful in a coilgun context.
If the coil is clamped with a standard rectifier diode there can still be a large
short term voltage transient at turn off due to the slow reverse recovery of the
diode. Normally a RC snubber or zener diode could be connected across the D-S
terminals in order to absorb the transient until the diode recovers. However these
are not needed if the transient energy is within the devices avalanche capacity.
A 'small' snubber may still be fitted to remove the ringing which occurs after
the diode begins conducting. Fig 3 shows the turn off transient current and voltage
(the test circuit used a small snubber).
3. Avalanche transient waveforms
we look at the D-S voltage we can see that it exceeds the rated 50V by about 20V
for a total duration of around 8us. This period represents the device under avalanche
breakdown. At the end of the breakdown the D-S voltage drops to just below 50V
at which point the commutating diode(s) takes over. During breakdown the current
through the D-S path decays and, interestingly, it goes negative before finally
returning to zero. I'm not quite sure why this happens but it could be due to
stray inductance in the current sensor - a redesign may be needed for measuring
microsecond events. The data sheets illustrate the current stopping when it first
reaches zero but it might due to either the commutating diode or mosfet body diode
recovery mechanisms. The negative current is not a consequence of the snubber
since it occurs with or without the snubber.
the D-S current and voltage are multiplied we can plot a power dissipation curve
for the breakdown. If this curve is integrated then the avalanche energy can be
calculated. Since I'm not sure why the current goes negative I'll ignore this
portion of the waveform and determine the energy dissipated up to this point.
Importing the data into Matlab and using the trapezium integration function yields
a value of 15.2mJ. This is energy is dissipated in only one device because it
is ver unlikely that the devices will have closely matched avalanche voltage levels.
Even so, this energy is within the 200mJ 'cold' junction avalanche capacity of
the BUZ11. Obviously the junction isn't going to be cold after conducting a large
current pulse and the better data sheets (many don't) provide enough information
on the avalanche characteristics to enable us to make an estimation of the allowable